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ISL6721
Data Sheet March 5, 2008 FN9110.6
Flexible Single-ended Current Mode PWM Controller
The ISL6721 is a low power, single-ended pulse width modulating (PWM) current mode controller designed for a wide range of DC/DC conversion applications including boost, flyback, and isolated output configurations. Peak current mode control effectively handles power transients and provides inherent overcurrent protection. Other features include a low power mode where the supply current drops to less than 200A during overvoltage and overcurrent shutdown faults. This advanced BiCMOS design features low operating current, adjustable operating frequency up to 1MHz, adjustable soft-start, and a bi-directional SYNC signal that allows the oscillator to be locked to an external clock for noise sensitive applications.
Features
* 1A MOSFET Gate Driver * 100A Startup Current * Fast Transient Response with Peak Current Mode Control * Adjustable Switching Frequency up to 1MHz * Bi-directional Synchronization * Low Power Disable Mode * Delayed Restart from OV and OC Shutdown Faults * Adjustable Slope Compensation * Adjustable Soft-start * Adjustable Overcurrent Shutdown Delay * Adjustable UV and OV Monitors * Leading Edge Blanking * Integrated Thermal Shutdown
Ordering Information
PART NUMBER ISL6721AB* PART MARKING ISL6721AB TEMP RANGE (C) PACKAGE PKG. DWG. # M16.15 M16.15
* 1% Tolerance Voltage Reference * Pb-Free Available (RoHS Compliant)
-40 to +105 16 Ld SOIC (150 mil) -40 to +105 16 Ld SOIC (150 mil) (Pb-Free) -40 to +105 16 Ld TSSOP (4.4mm)
ISL6721ABZ* 6721ABZ (Note) ISL6721AV* ISL67 21AV
Applications
* Telecom and Datacom Power * Wireless Base Station Power * File Server Power
M16.173 M16.173
ISL6721AVZ* ISL67 21AVZ -40 to +105 16 Ld TSSOP (Note) (4.4mm) (Pb-free)
* Industrial Power Systems * Isolated Buck and Flyback Regulators * Boost Regulators
*Add "-T" suffix for tape and reel. Please refer to TB347 for details on reel specifications. NOTE: These Intersil Pb-free plastic packaged products employ special Pb-free material sets; molding compounds/die attach materials and 100% matte tin plate PLUS ANNEAL - e3 termination finish, which is RoHS compliant and compatible with both SnPb and Pb-free soldering operations. Intersil Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020.
Pinout
ISL6721 (16 LD SOIC, TSSOP) TOP VIEW
GATE 1 ISENSE 2 SYNC 3 SLOPE 4 UV 5 OV 6 RTCT 7 ISET 8 16 VC 15 PGND 14 VCC 13 VREF 12 LGND 11 SS 10 COMP 9 FB
1
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures. 1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc. Copyright (c) Intersil Americas Inc. 2003-2005, 2007, 2008. All Rights Reserved. All other trademarks mentioned are the property of their respective owners.
ISL6721 Functional Block Diagram
VCC START/STOP UV COMPARATOR + BG + LGND THERMAL PROTECTION RESTART DELAY ISET ISENSE 5k VREF + S 53A + 100mV + + OC DETECT OVERCURRENT COMPARATOR Q Q 50s RETRIGGERABLE ONE SHOT SQ RQ OC LATCH 0.8 VREF 5V 1% VREF SOFT-START CHARGE 70A CURRENT ON SS CHARGE VOLTAGE CLAMP SS CHARGED
ENABLE
SS OVERCURRENT SHUTDOWN DELAY 25A + + 15A
4.375V ON
SLOPE 0.1
SS LOW
SS COMP + -
SS CLAMP PWM COMPARATOR ERROR AMPLIFIER + 1/3 + -
FAULT LATCH SQ RQ
+ SS LOW 270mV COMPARATOR + -
SET DOMINANT VREF UV COMPARATOR 4.65V + BG + -
VREF
2.5V VFB
VREF 20k 3.0V 1.5V 12k ON 30k OSCILLATOR COMPARATOR + 1mA VREF ON + + BLANKING COMPARATOR 3.0V + + SQ BI-DIRECTIONAL SYNCHRONIZATION OSC IN CLK OUT NO EXT SYNC EXT SYNC BLANKING SYNC IN SYNC OUT 36k RQ
2.50V + UV 1.45V + -
+ -
+ -
START 100ns BLANKING
+ -
OV
VC
RTCT
GATE
4V 2V
PGND
VREF
100k SYNC 4.5k
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FN9110.6 March 5, 2008
ISL6721 Typical Application - 48V Input Dual Output Flyback, 3.3V @ 2.5A, 1.8V @ 1.0A
SP1 SP2 CR5 T1 ISO LATIO N XF M R VIN+ P9 R21 +3.3V C21 + C15 + C16
R24
C18 CR4 C19 + C22 +
+1.8V
C2 C5
CR2
C17
C20 RET URN
CR6 R1 36-75V C6 C1 C3 TP1 Q1 R2 U2 C14 R16 R17 R18 R19
R4
R3
R22 U3 TP2
R15 C13
VIN-
R23
R20
R25 Q2 D1 TP3 SYNC C4 G AT E ISENSE SYNC U4 VC PG ND VCC
ISL6721
SLO PE UV R5 R6 D2 ISET TP5 OV RT CT
VREF L G ND SS R26 CO M P VFB R27
R14
T P4
Q3
C12 R8 R10 C11 C9 R11 R9 R12 R13 C10 C7 R7
VR1
C8
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FN9110.6 March 5, 2008
ISL6721 Typical Boost Converter Application Schematic
CR1 VIN+ L1 + C2 R12 C12 RETURN Q1 R8 R1 R2 R3 C11 C3 +VOUT
R4 C1
VIN+ C4 U1 VC GATE ISENSE PGND SYNC VCC SLOPE VREF UV LGND OV SS ISL6721 RTCT ISET R5 R11 C7 C9 VINC8 R7 R6 C6 COMP VFB R9 R10 C10
C5
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FN9110.6 March 5, 2008
ISL6721
Absolute Maximum Ratings
Supply Voltage, VCC, VC . . . . . . . . . . . . . . . . . GND -0.3V to +20.0V GATE . . . . . . . . . . . . . . . . GND - 0.3V to Gate Output Limit Voltage PGND to LGND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0.3V VREF . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to 5.3V Signal Pins . . . . . . . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to VREF Peak GATE Current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .1A
Thermal Information
Thermal Resistance (Typical, Note 1) JA (C/W) 16 Ld SOIC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 80 16 Ld TSSOP . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 105 Maximum Junction Temperature . . . . . . . . . . . . . . .-55C to +150C Maximum Storage Temperature Range . . . . . . . . . .-65C to +150C Pb-free reflow profile . . . . . . . . . . . . . . . . . . . . . . . . . .see link below http://www.intersil.com/pbfree/Pb-FreeReflow.asp
Operating Conditions
Temperature Range ISL6721Ax . . . . . . . . . . . . . . . . . . . . . . . . . . . .-40C to +105C Supply Voltage Range (Typical, Note 2) . . . . . . . . 9VDC to 18VDC
CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product reliability and result in failures not covered by warranty.
NOTES: 1. JA is measured with the component mounted on a high effective thermal conductivity test board in free air. See Tech Brief TB379 for details. 2. All voltages are with respect to GND.
Electrical Specifications
Recommended operating conditions unless otherwise noted. Refer to Block Diagram and Typical Application schematic on page 2 and page 3. 9V < VCC = VC < 20V, RT = 11k, Ct = 330 pF, TA = -40 to +105C (Note 3), Typical values are at TA = +25C. TEST CONDITIONS MIN TYP MAX UNITS
PARAMETER UNDERVOLTAGE LOCKOUT START Threshold STOP Threshold Hysteresis Start-Up Current, ICC OC/OV Fault Operating Current, ICC Operating Current, ICC Operating Supply Current, IC REFERENCE VOLTAGE Overall Accuracy
7.95 7.40 0.50 VCC < START Threshold Includes 1nF GATE loading -
8.25 7.70 0.55 100 200 4.5 8.0
8.55 8.20 1.00 175 300 10.0 12.0
V V V A A mA mA
Line, load, 0C to +105C Line, load, -40C to +105C
4.95 4.90 4.50 4.65 75 -10 -20
5.00 5.00 5 4.65 4.80 165 -
5.05 5.05 4.75 4.95 250 -
V V mV V V mV mA mA
Long Term Stability Fault Voltage VREF Good Voltage Hysteresis Operational Current Current Limit CURRENT SENSE Input Impedance Offset Voltage Input Voltage Range Blanking Time Gain, ACS
TA = +125C, 1000 hours (Note 5)
0.08 0 (Note 5) VSLOPE = 0V, VFB = 2.3V, VISET = 0.35V, 1.5V ACS = ISET/ISENSE 30 0.77
5 0.10 60 0.79
0.11 1.5 100 0.81
k V V ns V/V
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FN9110.6 March 5, 2008
ISL6721
Electrical Specifications
Recommended operating conditions unless otherwise noted. Refer to Block Diagram and Typical Application schematic on page 2 and page 3. 9V < VCC = VC < 20V, RT = 11k, Ct = 330 pF, TA = -40 to +105C (Note 3), Typical values are at TA = +25C. (Continued) TEST CONDITIONS MIN TYP MAX UNITS
PARAMETER ERROR AMPLIFIER Open Loop Voltage Gain Gain-Bandwidth Product Reference Voltage Initial Accuracy Reference Voltage COMP to PWM Gain, ACOMP COMP to PWM Offset FB Input Bias Current COMP Sink Current COMP Source Current COMP VOH COMP VOL PSRR SS Clamp, VCOMP OSCILLATOR Frequency Accuracy Frequency Variation with VCC Temperature Stability Maximum Duty Cycle Comparator High Threshold - Free Running Comparator High Threshold - with External SYNC Comparator Low Threshold Discharge Current SYNCHRONIZATION Input High Threshold Input Pulse Width Input Frequency Range Input Impedance VOH VOL SYNC Advance Output Pulse Width
(Note 5) (Note 5) VFB = COMP, TA = +25C (Note 5) VFB = COMP COMP = 4V, TA = +25C COMP = 4V (Note 5) VFB = 0V COMP = 1.5V, VFB = 2.7V COMP = 1.5V, VFB = 2.3V VFB = 2.3V VFB = 2.7V Frequency = 120Hz (Note 5) SS = 2.5V, VFB = 0V, ISET = 2V
60 2.465 2.44 0.31 0.51 -2 2 -0.25 4.25 0.4 60 2.4
90 15 2.515 2.515 0.33 0.75 0.1 6 -0.5 4.4 0.8 80 2.5
2.565 2.590 0.35 0.88 2 5.0 1.2 2.6
dB MHz V V V/V V A mA mA V V dB V
289 T = +105C (f20V - f9V)/f9V T = -40C (f20V -f9V)/f9V (Note 5) (Note 6) 68 (Note 5) 0C to +105C -40C to +105C 0.75 0.70
318 2 2 8 75 3.00 4.00 1.50 1.0 1.0
347 3 3 81 1.2 1.2
kHz % % % V V V mA
25 (Note 5) 0.65 x Free Running RLOAD = 4.5k RLOAD = open SYNC rising edge to GATE falling edge, CGATE = CSYNC = 100pF CSYNC = 100pF 2.5 50
4.5 25 -
2.5 1.0 0.1 55 -
V ns MHz k V V ns ns
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FN9110.6 March 5, 2008
ISL6721
Electrical Specifications
Recommended operating conditions unless otherwise noted. Refer to Block Diagram and Typical Application schematic on page 2 and page 3. 9V < VCC = VC < 20V, RT = 11k, Ct = 330 pF, TA = -40 to +105C (Note 3), Typical values are at TA = +25C. (Continued) TEST CONDITIONS MIN TYP MAX UNITS
PARAMETER SOFT-START Charging Current Charged Threshold Voltage Initial Overcurrent Discharge Current Overcurrent Shutdown Threshold Voltage Fault Discharge Current Reset Threshold Voltage SLOPE COMPENSATION Charge Current Slope Compensation Gain
SS = 2V
-40 4.26
-55 4.50 40 0.125 1.0 0.27
-70 4.74 55 0.155 0.31
A V A V mA V
Sustained OC Threshold < SS < Charged Threshold Charged Threshold minus, TA = +25C SS = 2V TA = +25C
30 0.095 0.25 0.22
SLOPE = 2V, 0C to +105C -40C to +105C Fraction of slope voltage added to ISENSE, TA = +25C Fraction of slope voltage added to ISENSE (Note 3)
-45 -41 0.097 0.082 -
-53 -53 0.1
-65 -65 0.103 0.118 0.2
A V/V V/V V
Discharge Voltage GATE OUTPUT Gate Output Limit Voltage Gate VOH Gate VOL Peak Output Current Output "Faulted" Leakage Rise Time Fall Time Minimum ON time
VRTCT = 4.5V
VC = 20V, CGATE = 1nF, IOUT = 0mA VC - GATE, VC = 10V, IOUT = 150mA GATE - PGND, IOUT = 150mA IOUT = 10mA VC = 20V, CGATE = 1nF (Note 5) VC = 20V, UV = 0V, GATE = 2V VC = 20V, CGATE = 1nF 1V < GATE < 9V VC = 20V, CGATE = 1nF 1V < GATE < 9V ISET = 0.5V; VFB = 0V; VC = 11V ISENSE to GATE w/10:1 Divider RTCT = 4.75V through 1k (Note 5)
11.0 1.2 -
13.5 1.5 1.2 0.6 1.0 2.6 60 15 -
16.0 2.2 1.5 0.8 100 40 110
V V V A mA ns ns ns
OVERCURRENT PROTECTION Minimum ISET Voltage Maximum ISET Voltage ISET Bias Current Restart Delay OV AND UV VOLTAGE MONITOR Overvoltage Threshold Undervoltage Fault Threshold Undervoltage Clear Threshold 2.4 1.38 1.41 2.5 1.45 1.53 2.6 1.52 1.62 V V V VISET = 1.00V TA = +25C 1.2 -1.0 150 295 0.35 1.0 445 V V A ms
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FN9110.6 March 5, 2008
ISL6721
Electrical Specifications
Recommended operating conditions unless otherwise noted. Refer to Block Diagram and Typical Application schematic on page 2 and page 3. 9V < VCC = VC < 20V, RT = 11k, Ct = 330 pF, TA = -40 to +105C (Note 3), Typical values are at TA = +25C. (Continued) TEST CONDITIONS MIN 20 VUV = 2.00 V VOV = 2.00 V -1.0 -1.0 TYP 50 MAX 100 1.0 1.0 UNITS mV A A
PARAMETER Undervoltage Hysteresis Voltage UV Bias Current OV Bias Current THERMAL PROTECTION Thermal Shutdown Thermal Shutdown Clear Hysteresis NOTES:
(Note 5) (Note 5) (Note 5)
120 105 -
130 120 10
140 135 -
C C C
3. Specifications at -40C and +105C are guaranteed by +25C test with margin limits. 4. This is the VCC current consumed when the device is active but not switching. Does not include gate drive current. 5. Limits should be considered typical and are not production tested. 6. This is the maximum duty cycle achievable using the specified values of RT and CT. Larger or smaller maximum duty cycles may be obtained using other values for RT and CT. See Equations 1, 2, 3 and 4.
Typical Performance Curves
NORMALIZED EA REFERENCE Normalized EA Reference 1.002 1.002 NORMALIZED Normalized VrefVREF 1.000 1 0.998 0.998 0.995 0.995 0.993 0.993 0.991 0.991 -40 1.002 1.002 1.000 1 0.998 0.998 0.995 0.995 0.993 0.993 0.991 0.991 -40
-10
20
50
80
110
-10
20
50
80
110
TEMPERATURE (C)
TEMPERATURE (C)
FIGURE 1. EA REFERENCE VOLTAGE vs TEMPERATURE
FIGURE 2. VREF REFERENCE VOLTAGE vs TEMPERATURE
1.002 NORMALIZED FREQUENCY 0.996 0.989 0.983 0.976 0.970 -40
103
FREQUENCY (kHz)
100pF 100 220pF 330pF 470pF 680pF 1000pF 2000pF 90 100
-10
20
50
80
110
10 10
20
30
40
TEMPERATURE (C)
50 60 70 RT (k)
80
FIGURE 3. OSCILLATOR FREQUENCY vs TEMPERATURE
FIGURE 4. RESISTANCE FOR CT CAPACITOR VALUES GIVEN
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FN9110.6 March 5, 2008
ISL6721 Pin Descriptions
SLOPE - Means by which the ISENSE ramp slope may be increased for improved noise immunity or improved control loop stability for duty cycles greater than 50%. An internal current source charges an external capacitor to GND during each switching cycle. The resulting ramp is scaled and added to the ISENSE signal. SYNC - A bidirectional synchronization signal used to coordinate the switching frequency of multiple units. Synchronization may be achieved by connecting the SYNC signal of each unit together or by using an external master clock signal. The oscillator timing capacitor, CT, is still required, even if an external clock is used. The first unit to assert this signal assumes control. RTCT - This is the oscillator timing control pin. The operational frequency and maximum duty cycle are set by connecting a resistor, RT, between VREF and this pin and a timing capacitor, CT, from this pin to LGND. The oscillator produces a sawtooth waveform with a programmable frequency range of 100kHz to 1.0MHz. The charge time, tC, the discharge time, tD, the switching frequency, fsw, and the maximum duty cycle, Dmax, can be calculated from Equations 1, 2, 3 and 4:
t C 0.655 * R T * C T S (EQ. 1)
UV - Undervoltage monitor input pin. This signal is compared to an internal 1.45V reference to detect an undervoltage condition. ISENSE - This is the input to the current sense comparators. The IC has two current sensing comparators, a PWM comparator for peak current mode control, and an overcurrent protection comparator. The overcurrent comparator threshold is adjustable through the ISET pin. Exceeding the overcurrent threshold will start a delayed shutdown sequence. Once an overcurrent condition is detected, the soft-start charge current source is disabled and a discharge current source is enabled. The soft-start capacitor begins discharging, and if it discharges to less than 4.375V (sustained overcurrent threshold), a shutdown condition occurs and the GATE output is forced low. At this point a reduced discharge current takes over until the soft-start voltage reaches 0.27V (reset threshold). The GATE output remains low until the reset threshold is attained. At this point, a soft-start cycle begins. If the overcurrent condition ceases, and then an additional 50s period elapses before the shutdown threshold is reached, no shutdown occurs and the soft-start voltage is allowed to recharge. LGND - LGND is a small signal reference ground for all analog functions on this device. PGND - This pin provides a dedicated ground for the output gate driver. The LGND and PGND pins should be connected externally using a short printed circuit board trace close to the IC. This is imperative to prevent large, high frequency switching currents flowing through the ground metallization inside the IC. (Decouple VC to PGND with a low ESR 0.1F or larger capacitor.) GATE - This is the device output. It is a high current power driver capable of driving the gate of a power MOSFET with peak currents of 1.0A. This GATE output is actively held low when VCC is below the UVLO threshold. The output high voltage is clamped to ~13.5V. Voltages exceeding this clamp value should not be applied to the GATE pin. The output stage provides very low impedance to overshoot and undershoot. VC - This pin is for separate collector supply to the output gate drive. Separate VC and PGND helps decouple the IC's analog circuitry from the high power gate drive noise. (Decouple VC to PGND with a low ESR 0.1F or larger capacitor.) VCC - VCC is the power connection for the device. Although quiescent current, ICC, is low, it is dependent on the frequency of operation. To optimize noise immunity, bypass VCC to LGND with a ceramic capacitor as close to the VCC and LGND pins as possible.
0.001 * R T - 3.6 t D - R * C * LN ------------------------------------------ T T 0.001 * R T - 1.9
S
(EQ. 2)
1 f sw = ---------------tD + tC
Hz
(EQ. 3)
Dmax = t C * f sw
(EQ. 4)
Figure 4 may be used as a guideline in selecting the capacitor and resistor values required for a given frequency. COMP - COMP is the output of the error amplifier and the input of the PWM comparator. The control loop frequency compensation network is connected between the COMP and FB pins. The ISL6721 features a built-in full cycle soft-start. Soft-start is implemented as a clamp on the maximum COMP voltage. FB - Feedback voltage input connected to the inverting input of the error amplifier. The non-inverting input of the error amplifier is internally tied to a reference voltage. Current sense leading edge blanking is disabled when the FB input is less than 2.0V. OV - Overvoltage monitor input pin. This signal is compared to an internal 2.5V reference to detect an overvoltage condition.
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ISL6721
The total supply current (IC plus ICC) will be higher, depending on the load applied to GATE. Total current is the sum of the quiescent current and the average gate current. Knowing the operating frequency, fsw, and the MOSFET gate charge, Qg, the average GATE output current can be calculated in Equation 5:
Igate = Qg * f sw A (EQ. 5)
VREF - The 5V reference voltage output. Bypass to LGND with a 0.01F or larger capacitor to filter this output as needed. Using capacitance less than this value may result in unstable operation. SS - Connect the soft-start capacitor between this pin and LGND to control the duration of soft-start. The value of the capacitor determines both the rate of increase of the duty cycle during start-up, and also controls the overcurrent shutdown delay. ISET - A DC voltage between 0.35V and 1.2V applied to this input sets the pulse-by-pulse overcurrent threshold. When overcurrent inception occurs, the SS capacitor begins to discharge and starts the overcurrent delayed shutdown cycle.
During normal operation the RTCT voltage charges from 1.5V to 3.0V and back during each cycle. Clock and SYNC signals are generated when the 3.0V threshold is reached. If an external clock signal is detected during the latter 2/3 of the charging cycle, the oscillator switches to external synchronization mode and relies upon the external SYNC signal to terminate the oscillator cycle. The generation of a SYNC signal is inhibited in this mode. If the RTCT voltage exceeds 4.0V (i.e. no external SYNC signal terminates the cycle), the oscillator reverts to the internal clock mode and a SYNC signal is generated.
Soft-Start Operation
The ISL6721 features soft-start using an external capacitor in conjunction with an internal current source. Soft-start is used to reduce voltage stresses and surge currents during start up. Upon start up, the soft-start circuitry clamps the error amplifier output (COMP pin) to a value proportional to the soft-start voltage. The error amplifier output rises as the soft-start capacitor voltage rises. This has the effect of increasing the output pulse width from zero to the steady state operating duty cycle during the soft-start period. When the soft-start voltage exceeds the error amplifier voltage, soft-start is completed. Soft-start forces a controlled output voltage rise. Soft-start occurs during start-up and after recovery from a fault condition or overcurrent shutdown. The soft-start voltage is clamped to 4.5V.
Functional Description
Features
The ISL6721 current mode PWMs make an ideal choice for low-cost flyback and forward topology applications requiring enhanced control and supervisory capability. With adjustable overvoltage and undervoltage thresholds, overcurrent threshold, and hic-cup delay, a highly flexible design with minimal external components is possible. Other features include peak current mode control, adjustable soft-start, slope compensation, adjustable oscillator frequency, and a bi-directional synchronization clock input.
Gate Drive
The ISL6721 is capable of sourcing and sinking 1A peak current. Separate collector supply (VC) and power ground (PGnd) pins help isolate the IC's analog circuitry from the high power gate drive noise. To limit the peak current through the IC, an external resistor may be placed between the totem-pole output of the IC (GATE pin) and the gate of the MOSFET. This small series resistor also damps any oscillations caused by the resonant tank of the parasitic inductances in the traces of the board and the FET's input capacitance.
Oscillator
The ISL6721 have a sawtooth oscillator with a programmable frequency range to 1MHz, which can be programmed with a resistor and capacitor on the RTCT pin. (Please refer to Figure 4 for the resistance and capacitance required for a given frequency.)
Slope Compensation
For applications where the maximum duty cycle is less than 50%, slope compensation may be used to improve noise immunity, particularly at lighter loads. The amount of slope compensation required for noise immunity is determined empirically, but is generally about 10% of the full scale current feedback signal. For applications where the duty cycle is greater than 50%, slope compensation is required to prevent instability. Slope compensation is a technique in which the current feedback signal is modified by adding additional slope to it. The minimum amount of slope compensation required corresponds to 1/2 the inductor downslope. However, adding excessive slope compensation results in a control loop that behaves more as a voltage mode controller than as current mode controller.
Implementing Synchronization
The oscillator can be synchronized to an external clock applied at the SYNC pin or by connecting the SYNC pins of multiple ICs together. If an external master clock signal is used, it must be at least 65% of the free running frequency of the oscillator for proper synchronization. The external master clock signal should have a pulse width greater than 20ns. If no master clock is used, the first device to assert SYNC assumes control of the SYNC signal. An external SYNC pulse is ignored if it occurs during the first 1/3 of the switching cycle.
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FN9110.6 March 5, 2008
ISL6721
DOWNSLOPE Downslope
ISENSE SIGNAL (V)
CURRENT SENSE SIGNAL Current Sense Signal
A resistor divider between VIN and LGND to each input determines the operational thresholds. The UV threshold has a fixed hysteresis of 75mV nominal.
Overcurrent Operation
The overcurrent threshold level is set by the voltage applied at the ISET pin. Setting the overcurrent level may be accomplished by using a resistor divider network from VREF to LGND. The ISET threshold should be set at a level that corresponds to the desired peak output inductor current plus the additive effects of slope compensation. Overcurrent delayed shutdown is enabled once the soft-start cycle is complete. If an overcurrent condition is detected, the soft-start charging current source is disabled and the discharging current source is enabled. The soft-start capacitor is discharged at a rate of 40A. At the same time, a 50s retriggerable one-shot timer is activated amd it remains active for 50s after the overcurrent condition stops. The soft-start discharge cycle cannot be reset until the oneshot timer becomes inactive. If the soft-start capacitor discharges by more than 0.125V to 4.375V, the output is disabled and the soft-start capacitor is discharged. The output remains disabled and ICC drops to 200A for approximately 295ms. A new soft-start cycle is then initiated. The shutdown and restart behavior of the OC protection is often referred to as hic-cup operation due to its repetitive start-up and shutdown characteristic. If the overcurrent condition ceases at least 50s prior to the soft-start voltage reaching 4.375V, the soft-start charging and discharging currents revert to normal operation and the soft-start voltage is allowed to recover. Hiccup OC protection may be defeated by setting ISET to a voltage that exceeds the Error Amplifier current control voltage, or about 1.5V.
TIME Time
FIGURE 5.
The minimum amount of capacitance to place at the SLOPE pin is calculated in Equation 6:
C SLOPE = 4.24 x10
-6
t ON * ---------------------V SLOPE
F
(EQ. 6)
where tON is the On time and VSLOPE is the amount of voltage to be added as slope compensation to the current feedback signal. In general, the amount of slope compensation added is 2 to 3 times the minimum required. Example: Assume the inductor current signal presented at the ISENSE pin decreases 125mV during the Off period, and: Switching Frequency, fsw = 250kHz Duty Cycle, D = 60% tON = D/fsw = 0.6/250E3 = 2.4s tOFF = (1 - D)/fsw = 1.6s Determine the downslope: Downslope = 0.125V/1.6s = 78mV/s. Now determine the amount of voltage that must be added to the current sense signal by the end of the On time.
1 V SLOPE = -- * 0.078 * 2.4 = 94mV 2 (EQ. 7)
Leading Edge Blanking
-6
Therefore,
C SLOPE ( MIN ) = 4.24 x10
-6
2.4 x10 * ----------------------- 110pF 0.094
(EQ. 8)
An appropriate slope compensation capacitance for this example would be 1/2 to 1/3 the calculated value, or between 68pF and 33pF.
The initial 100ns of the current feedback signal input at ISENSE is removed by the leading edge blanking circuitry. The blanking period begins when the GATE output leading edge exceeds 3.0V. Leading edge blanking prevents current spikes from parasitic elements in the power supply from causing false trips of the PWM comparator and the overcurrent comparator.
Overvoltage and Undervoltage Monitor
The OV and UV signals are inputs to a window comparator used to monitor the input voltage level to the converter. If the voltage falls outside of the user designated operating range, a shutdown fault occurs. For OV faults, the supply current, ICC, is reduced to 200A for ~295ms at which time recovery is attempted. If the fault is cleared, a soft-start cycle begins. Otherwise another shutdown cycle occurs. A UV condition also results in a shutdown fault, but the device does not enter the low power mode and no restart delay occurs when the fault clears. 11
Fault Conditions
A Fault condition occurs if VREF falls below 4.65V, the OV input exceeds 2.50V, the UV input falls below 1.45V, or the junction temperature of the die exceeds ~+130C. When a Fault is detected the GATE output is disabled and the soft-start capacitor is quickly discharged. When the Fault condition clears and the soft-start voltage is below the reset threshold, a soft-start cycle begins.
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ISL6721
Ground Plane Requirements
Careful layout is essential for satisfactory operation of the device. A good ground plane must be employed. A unique section of the ground plane must be designated for high di/dt currents associated with the output stage. Power ground (PGND) can be separated from the logic ground (LGND) and connected at a single point. VC should be bypassed directly to PGND with good high frequency capacitors. The return connection for input power and the bulk input capacitor should be connected to the PGND ground plane. POUT: 10W Efficiency: 70% Maximum Duty Cycle, DMAX: 0.45
Transformer Design
The design of a flyback transformer is a non-trivial affair. It is an iterative process which requires a great deal of experience to achieve the desired result. It is a process of many compromises, and even experienced designers will produce different designs when presented with identical requirements. The iterative design process is not presented here for clarity. The abbreviated design process follows: * Select a core geometry suitable for the application. Constraints of height, footprint, mounting preference, and operating environment will affect the choice. * Select suitable core material(s). * Select maximum flux density desired for operation. * Select core size. Core size will be dictated by the capability of the core structure to store the required energy, the number of turns that have to be wound, and the wire gauge needed. Often the window area (the space used for the windings) and power loss determine the final core size. For flyback transformers, the ability to store energy is the critical factor in determining the core size. The cross sectional area of the core and the length of the air gap in the magnetic path determine the energy storage capability. * Determine maximum desired flux density. Depending on the frequency of operation, the core material selected, and the operating environment, the allowed flux density must be determined. The decision of what flux density to allow is often difficult to determine initially. Usually the highest flux density that produces an acceptable design is used, but often the winding geometry dictates a larger core than is required based on flux density and energy storage calculations. * Determine the number of primary turns. * Determine the turns ratio. * Select the wire gauge for each winding. * Determine winding order and insulation requirements. * Verify the design. Input Power: POUT/Efficiency = 14.3W (use 15W) Max On Time: tON(MAX) = DMAX/fsw = 2.25s Average Input Current: IAVG(IN) = PIN/VIN(MIN) = 0.42A Peak Primary Current:
2 * I AVG ( IN ) I PPK = ---------------------------------------- = 1.87 f sw * t ON ( MAX ) A (EQ. 9)
Reference Design
The Typical Application Schematic on page 3 features the ISL6721 in a conventional dual output 10W discontinuous mode flyback DC/DC converter. The ISL6721EVAL1 demonstration unit implements this design and is available for evaluation. The input voltage range is from 36VDC to 75VDC, and the two outputs are 3.3V @ 2.5A and 1.8V @ 1.0A. Cross regulation is achieved using the weighted sum of the two outputs.
Circuit Element Descriptions
The converter design may be broken down into the following functional blocks: Input Storage and Filtering Capacitance: C1, C2, C3 Isolation Transformer: T1 Primary voltage Clamp: CR6, R24, C18 Start Bias Regulator: R1, R2, R6, Q3, VR1 Operating Bias and Regulator: R25, Q2, D1, C5, CR2, D2 Main MOSFET Power Switch: Q1 Current Sense Network: R4, R3, R23, C4 Feedback Network:, R13, R15, R16, R17, R18, R19, R20, R26, R27, C13, C14, U2, U3 Control Circuit:C7, C8, C9, C10, C11, C12, R5, R6, R8, R9, R10, R11, R12, R14, R22 Output Rectification and Filtering: CR4, CR5, C15, C16, C19, C20, C21, C22 Secondary Snubber: R21, C17
Design Criteria
The following design requirements were selected: Switching Frequency, fsw: 200kHz VIN: 36V to 75V VOUT(1): 3.3V @ 2.5A VOUT(2): 1.8V @ 1.0A VOUT(BIAS): 12V @ 50mA
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Maximum Primary Inductance:
V IN ( MIN ) * t ON ( MAX ) Lp ( max ) = -------------------------------------------------------- = 43.3 I PPK H (EQ. 10)
Since:
o * N p * Aeff L p = ---------------------------------------lg
2
H
(EQ. 13)
Choose desired primary inductance to be 40H. The core structure must be able to deliver a certain amount of energy to the secondary on each switching cycle in order to maintain the specified output power.
V OUT + Vd w = P OUT * ----------------------------------f sw * V OUT joules (EQ. 11)
the number of primary turns, Np, may be calculated. The result is Np = 40 turns. The secondary turns may be calculated as follows:
Ig * Vout + Vd * tr N s ------------------------------------------------------N p * Ippk * o * Aeff (EQ. 14)
where w is the amount of energy required to be transferred each cycle and Vd is the drop across the output rectifier. The capacity of a gapped ferrite core structure to store energy is dependent on the volume of the airgap and can be expressed in Equation 12:
2 * o * w Vg = Aeff * lg = ----------------------------2 B m
3
(EQ. 12)
where Aeff is the effective cross sectional area of the core in m2, lg is the length of the airgap in meters, o is the permeability of free space (4 * 10-7), and B is the change in flux density in Tesla. A core structure having less airgap volume than calculated will be incapable of providing the full output power over some portion of its operating range. On the other hand, if the length of the airgap becomes large, magnetic field fringing around the gap occurs. This has the effect of increasing the airgap volume. Some fringing is usually acceptable, but excessive fringing can cause increased losses in the windings around the gap resulting in excessive heating. Once a suitable core and gap combination are found, the iterative design cycle begins. A design is developed and checked for ease of assembly and thermal performance. If the core does not allow adequate space for the windings, then a core with a larger window area is required. If the transformer runs hot, it may be necessary to lower the flux density (more primary turns, lower operating frequency), select a less lossy core material, change the geometry of the windings (winding order), use heavier gauge wire or multi-filar windings, and/or change the type of wire used (Litz wire, for example). For simplicity, only the final design is further described. An EPCOS EFD 20/10/7 core using N87 material gapped to an AL value of 25nH/N2 was chosen. It has more than the required air gap volume to store the energy required, but was needed for the window area it provides. Aeff = 31 * 10-6 lg = 1.56 * 10-3 m2 m
where tr is the time required to reset the core. Since discontinuous MMF mode operation is desired, the core must completely reset during the off time. To maintain discontinuous mode operation, the maximum time allowed to reset the core is tsw - tON(MAX) where tsw = 1/fsw. The minimum time is application dependent and at the designers discretion knowing that the secondary winding RMS current and ripple current stress in the output capacitors increases with decreasing reset time. The calculation for maximum Ns for the 3.3 V output using t = tsw - tON (MAX) = 2.75s is 5.52 turns. The determination of the number of secondary turns is also dependent on the number of outputs and the required turns ratios required to generate them. If Schottky output rectifiers are used and we assume a forward voltage drop of 0.45V, the required turns ratio for the two output voltages, 3.3V and 1.8V, is 5:3. With a turns ratio of 5:3 for the secondary windings, we will use Ns1 = 5 turns and Ns2 = 3 turns. Checking the reset time using these values for the number of secondary turns yields a duration of Tr = 2.33s or about 47% of the switching period, an acceptable result. The bias winding turns may be calculated similarly, only a diode forward drop of 0.7V is used. The rounded off result is 17 turns for a 12V bias. The next step is to determine the wire gauge. The RMS current in the primary winding may be calculated using Equation 15:
t ON ( MAX ) I P ( RMS ) = I PPK * -------------------------3 * t sw A (EQ. 15)
The peak and RMS current values in the remaining windings may be calculated using Equations 16 and 17:
2 * I OUT * t sw I SPK = -----------------------------------Tr A (EQ. 16)
t sw I RMS = 2 * I OUT * -------------3 * Tr
A
(EQ. 17)
The flux density B is only 0.069T or 690 gauss, a relatively low value.
The RMS current for the primary winding is 0.72A, for the 3.3V output, 4.23A, for the 1.8V output, 1.69A, and for the bias winding, 85mA.
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To minimize the transformer leakage inductance, the primary was split into two sections connected in parallel and positioned such that the other windings were sandwiched between them. The output windings were configured so that the 1.8V winding is a tap off of the 3.3V winding. Tapping the 1.8V output requires that the shared portion of the secondary conduct the combined current of both outputs. The secondary wire gauge must be selected accordingly. The determination of current carrying capacity of wire is a compromise between performance, size, and cost. It is affected by many design constraints such as operating frequency (harmonic content of the waveform) and the winding proximity/geometry. It generally ranges between 250 and 1000 circular mils per ampere. A circular mil is defined as the area of a circle 0.001" (1 mil) in diameter. As the frequency of operation increases, the AC resistance of the wire increases due to skin and proximity effects. Using heavier gauge wire may not alleviate the problem. Instead multiple strands of wire in parallel must be used. In some cases, Litz wire is required. The winding configuration selected is: Primary #1: 40T, 2 #30 bifilar Secondary: 5T, 0.003" (3 mil) copper foil tapped at 3T Bias: 17T #32 Primary #2: 40T, 2 #30 bifilar The internal spacing and insulation system was designed for 1500VDC dielectric withstand rating between the primary and secondary windings. device to enter a thermal runaway situation without proper heatsinking. As a general rule of thumb, doubling the +25C rDS(ON) specification yields a reasonable value for estimating the conduction losses at +125C junction temperature. The switching losses have two components, capacitive switching losses and voltage/current overlap losses. The capacitive losses occur during turn on of the device and may be calculated in Equation 19:
2 1 Pswcap = -- * Cfet * Vin * f sw 2
W
(EQ. 19)
where Cfet is the equivalent output capacitance of the MOSFET. Device output capacitance is specified on datasheets as Coss and is non-linear with applied voltage. To find the equivalent discrete capacitance, Cfet, a charge model is used. Using a known current source, the time required to charge the MOSFET drain to the desired operating voltage is determined and the equivalent capacitance may be calculated in Equation 20:
Ichg * t Cfet = ------------------V F (EQ. 20)
The other component of the switching loss is due to the overlap of voltage and current during the switching transition. A switching transition occurs when the MOSFET is in the process of either turning on or off. Since the load is inductive, there is no overlap of voltage and current during the turn on transition, so only the turn off transition is of significance. The power dissipation may be estimated using Equation 21:
1 P sw -- * I PPK * V IN * t OL * f sw x (EQ. 21)
Power MOSFET Selection
Selection of the main switching MOSFET requires consideration of the voltage and current stresses that will be encountered in the application, the power dissipated by the device, its size, and its cost. The input voltage range of the converter is 36VDC to 75VDC. This suggests a MOSFET with a voltage rating of 150V is required due to the flyback voltage likely to be seen on the primary of the isolation transformer. The losses associated with MOSFET operation may be divided into three categories: conduction, switching, and gate drive. The conduction losses are due to the MOSFET's ON resistance.
Pcond = r DS ( ON ) * Iprms
2
where tOL is the duration of the overlap period and x ranges from about 3 through 6 in typical applications and depends on where the waveforms intersect. This estimate may predict higher dissipation than is realized because a portion of the turn off drain current is attributable to the charging of the device output capacitance (Coss) and is not dissipative during this portion of the switching cycle.
Ip p k
W
(EQ. 18)
V D -S Tol
where rDS(ON) is the ON resistance of the MOSFET and Iprms is the RMS primary current. Determining the conduction losses is complicated by the variation of rDS(ON) with temperature. As junction temperature increases, so does rDS(ON), which increases losses and raises the junction temperature more, and so on. It is possible for the 14
FIGURE 6. SWITCHING CYCLE
The final component of MOSFET loss is caused by the charging of the gate capacitance through the device gate resistance. Depending on the relative value of any external
FN9110.6 March 5, 2008
ISL6721
resistance in the gate drive circuit, a portion of this power will be dissipated externally.
Pgate = Qg * Vg * f sw W (EQ. 22) ( Ispk - Iout ) * Tr ( 10.73 - 2.5 ) * 2.33 x10 C --------------------------------------------- = ------------------------------------------------------------------ = 960F 2 * 0.010 2 * V (EQ. 24)
-6
Once the losses are known, the device package must be selected and the heatsinking method designed. Since the design requires a small surface mount part, a 8 Ld SOIC package was selected. A Fairchild FDS2570 MOSFET was selected based on these criteria. The overall losses are estimated at 400mW.
ESL adds to the ripple and noise voltage in proportion to the rate of change of current into the capacitor (V = L * di/dt).
V * dt 0.030 * 200 x10 L -------------- = --------------------------------------------- = 0.56nH 10.73 di
-9
(EQ. 25)
Output Filter Design
In a flyback design, the primary concern for the design of the output filter is the capacitor ripple current stress and the ripple and noise specification of the output. The current flowing in and out of the output capacitors is the difference between the winding current and the output current. The peak secondary current, ISPK, is 10.73A for the 3.3V output and 4.29A for the 1.8V output. The current flowing into the output filter capacitor is the difference between the winding current and the output current. Looking at the 3.3V output, the peak winding current is ISPK = 10.73A. The capacitor must store this amount minus the output current of 2.5A, or 8.23A. The RMS ripple current in the 3.3V output capacitor is about 3.5ARMS. The RMS ripple current in the 1.8V output capacitor is about 1.4ARMS. Voltage deviation on the output during the switching cycle (ripple and noise) is caused by the change in charge of the output capacitance, the equivalent series resistance (ESR), and equivalent series inductance (ESL). Each of these components must be assigned a portion of the total ripple and noise specification. How much to allow for each contributor is dependent on the capacitor technology used. For purposes of this discussion, we will assume the following: 3.3V output: 100mV total output ripple and noise ESR: 60mV Capacitor Q: 10mV ESL: 30mV 1.8V output: 50mV total output ripple and noise ESR: 30mV Capacitor Q: 5mV
Capacitors having high capacitance usually do not have sufficiently low ESL. High frequency capacitors such as surface mount ceramic or film are connected in parallel with the high capacitance capacitors to address the effects of ESL. A combination of high frequency and high ripple capability capacitors is used to achieve the desired overall performance. The analysis of the 1.8V output is similar to that of the 3.3V output and is omitted for brevity. Two OSCON 4SEP560M (560F) electrolytic capacitors and a 22F X5R ceramic 1210 capacitor were selected for both the 3.3 and 1.8V outputs. The 4SEP560M electrolytic capacitors are each rated at 4520mA ripple current and 13m of ESR. The ripple current rating of just one of these capacitors is adequate, but two are needed to meet the minimum ESR and capacitance values. The bias output is of such low power and current that it places negligible stress on its filter capacitor. A single 0.1F ceramic capacitor was selected.
Control Loop Design
The major components of the feedback control loop are a programmable shunt regulator, an opto-coupler, and the inverting amplifier of the ISL6721. The opto-coupler is used to transfer the error signal across the isolation barrier. The opto-coupler offers a convenient means to cross the isolation barrier, but it adds complexity to the feedback control loop. It adds a pole at about 10kHz and a significant amount of gain variation due the current transfer ratio (CTR). The CTR of the opto-coupler varies with initial tolerance, temperature, forward current, and age. A block diagram of the feedback control loop is shown in Figure 7.
PRIMARY SIDE AMPLIFIER REF + PWM Z3 Z4
POWER STAGE
VOUT
ESL: 15mV For the 3.3V output:
V 0.060 ESR -------------------------------- = ---------------------------- = 7.3m I SPK - I OUT 10.73 - 2.5 (EQ. 23)
ISOLATION
ERROR AMPLIFIER Z2 + REF Z1
The change in voltage due to the change in charge of the output capacitor, Q, determines how much capacitance is required on the output.
FIGURE 7. FEEDBACK CONTROL LOOP
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ISL6721
The loop compensation is placed around the Error Amplifier (EA) on the secondary side of the converter. The primary side amplifier located in the control IC is used as a unity gain inverting amplifier and provides no loop compensation. A Type 2 error amplifier configuration was selected as a precaution in case operation in continuous mode should occur at some operating point.
VOUT
I spk ( max ) K = ------------------------V c ( max ) R o = LoadResis tan ce L s = SecondaryInduc tan ce 2 p = ------------------Ro * Co 1 z = ------------------Rc * Co or or 1 f p = ---------------------------- * Ro * Co 1 f z = ------------------------------------2 * * Rc * Co
(EQ. 28)
(EQ. 29) (EQ. 30) (EQ. 31) (EQ. 32) (EQ. 33) (EQ. 34) (EQ. 35)
VERROR + REF
C o = OutputCapaci tan ce R c = OutputCapaci tan ceE SR V c ( max ) = ControlVoltageRange
FIGURE 8. TYPE 2 ERROR AMPLIFIER
Development of a small signal model for current mode control is rather complex. The method of reference1 was selected for its ability to accurately predict loop behavior. To further simplify the analysis, the converter will be modeled as a single output supply with all of the output capacitance reflected to the 3.3V output. Once the "single" output system is compensated, adjustments to the compensation will be required based on actual loop measurements. The first parameter to determine is the peak current feedback loop gain. Since this application is low power, a resistor in series with the source of the power switching MOSFET is used for the current feedback signal. For higher power applications, a resistor would dissipate too much power and current transformer would be used instead. There is limited flexibility to adjust the current loop behavior due to the need to provide overcurrent protection. Current limit and the current loop gain are determined by the current sense resistor and the ISET threshold. ISET was set at 1.0V, near its maximum, to minimize noise effects. When determining ISET, the internal gain and offset of the ISENSE signal in the control IC must be taken into account. The maximum peak primary current was determined earlier to be 1.87A, so a choice of 2.25A peak primary current for current limit is reasonable. A current gain, AEXT, of 0.5V/A was selected to achieve this.
ISET = 2.25 * 0.8 * 0.5 + 0.100 = 1.00 V (EQ. 26)
The value of K may be determined by assuming all of the output power is delivered by the 3.3V output at the threshold of current limit. The maximum power allowed was determined earlier as 15W, therefore:
P out -6 152 * ----------- * t sw 2 * ------- * 5 x10 V out 3.3 I spk ( max ) = ----------------------------------- = ----------------------------------------- = 19.5 -6 Tr 2.33 x10 1 v c ( max ) = V ISENSE * A EXT * A CS * -------------------- = 2.93 A COMP
A (EQ. 36) V (EQ. 37)
where AEXT is the external gain of the current feedback network, ACS is the IC internal gain, and ACOMP is the gain between the error amplifier and the PWM comparator. The Type 2 compensation configuration has two poles and one zero. The first pole is at the origin, and provides the integration characteristic which results in excellent DC regulation. Referring to the Typical Application Schematic on page 3, the remaining pole and zero for the compensator are located at:
C 13 + C 14 1 f pc = ------------------------------------------------------------ ------------------------------------------2 * * R 15 * C 14 * C 13 2 * * R 15 * C 14 1 f zc = ------------------------------------------2 * * R 15 * C 13 (EQ. 38)
(EQ. 39)
The ratio of R15 to the parallel combination of R17 and R18 determine the mid band gain of the error amplifier.
R 15 * ( R 17 + R 18 ) A midband = ----------------------------------------------R 17 * R 18 (EQ. 40)
The control to output transfer function may be represented as2:
s 1 + -----vo z R o * L s * f sw ----- = K * --------------------------------- * ---------------vc s 2 1 + -----p
(EQ. 27)
If we ignore the current feedback sampled-data effects:
From Equation 27, it can be seen that the control to output transfer function frequency dependence is a function of the output load resistance, the value of output capacitance, and the output capacitance ESR. These variations must be considered when compensating the control loop. The worst case small signal operating point for the converter is at
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minimum VIN, maximum load, maximum COUT, and minimum ESR. The higher the desired bandwidth of the converter, the more difficult it is to create a solution that is stable over the entire operating range. A good rule of thumb is to limit the bandwidth to about fsw/4. For this example, the bandwidth will be further limited due to the low GBWP of the LM431-based Error Amplifier and the opto-coupler. A bandwidth of approximately 5kHz was selected. For the EA compensation, the first pole is placed at the origin by default (C14 is an integrating capacitor). The first zero is placed below the crossover frequency, fco, usually around 1/3 fco. The second pole is placed at the lower of the ESR zero or at one half of the switching frequency. The midband gain is then adjusted to obtain the desired crossover frequency. If the phase margin is not adequate, the crossover frequency may have to be reduced. Using this technique to determine the compensation, the following values for the EA components were selected. R17 = R18 = R15 = 1k R20 = open C13 = 100nF C14 = 100pF A Bode plot of the closed loop system at low line, max load appears in Figures 9A and 9B.
50 40 30 20 10 0 -10 -20 -30 -40 -50 10k
Regulation Performance
TABLE 1. OUTPUT LOAD REGULATION, VIN = 48V IOUT (A), 3.3V 0 0.39 0.88 1.38 1.87 2.39 2.89 3.37 0 0.39 0.88 1.38 1.87 2.39 2.89 0 0.39 0.88 1.38 1.87 2.39 0 0.39 0.88 1.38 1.87 0 0.39 0.88 IOUT (A), 1.8V VOUT (V), 3.3V VOUT (V), 1.8V 0.030 0.030 0.030 0.030 0.030 0.030 0030 0.030 0.52 0.52 0.52 0.52 0.52 0.52 0.52 1.05 1.05 1.05 1.05 1.05 1.05 1.55 1.55 1.55 1.55 1.55 2.07 2.07 2.07 2.07 2.62 2.62 2.62 3.14 3.14 3.351 3.281 3.251 3.223 3.204 3.185 3.168 3.153 3.471 3.283 3.254 3.233 3.218 3.203 3.191 3.619 3.290 3.254 3.235 3.220 3.207 3.699 3.306 3.260 3.239 3.224 3.762 3.329 3.270 3.245 3.819 3.355 3.282 3.869 3.383 1.825 1.956 1.988 2.014 2.029 2.057 2.084 2.103 1.497 1.800 1.836 1.848 1.855 1.859 1.862 1.347 1.730 1.785 1.805 1.814 1.820 1.265 1.682 1.750 1.776 1.789 1.201 1.645 1.722 1.752 1.142 1.612 1.697 1.091 1.581
GAIN (dB)
100k
1M
10M
100M
1.38 0 0.39 0.88
FREQUENCY (Hz)
FIGURE 9A. GAIN
200 PHASE MARGIN () 150 100 50 0 -50 -100 10k 100k 1M 10M 100M
0 0.39
Waveforms
Typical waveforms can be found in Figures 10 through 12. Figure 10 shows the steady state operation of the sawtooth oscillator waveform at RTCT (Trace 2), the SYNC output pulse (Trace 1), and the GATE output to the converter FET (Trace 3). Figure 11 shows the converter behavior while operating in an overcurrent fault condition. Trace 1 is the soft-start voltage, which increases from 0V to 4.5V, at which point the OC fault function is enabled. The OC condition is detected and the soft-start capacitor is discharged to the
FN9110.6 March 5, 2008
FREQUENCY (Hz)
FIGURE 9B. PHASE MARGIN
17
ISL6721
4.375V OC fault threshold at which point the IC enters the fault shutdown mode. Trace 2 shows the behavior of the timing capacitor voltage during a shutdown fault. Most of the functions of the IC are de-powered during a fault, and the oscillator is among those functions. During a fault, the IC is turned off until the restart delay has timed out. After the delay, power is restored and the IC resumes normal operation. Trace 3 is the GATE output during the soft-start cycle and OC fault.
NOTE: Trace 1: VD-S Trace 3: VG-S FIGURE 12. GATE AND DRAIN-SOURCE WAVEFORMS
NOTE: Trace 1: SYNC Output Trace 2: RTCT Sawtooth Trace 3: GATE Output FIGURE 10. TYPICAL WAVEFORMS
NOTE: Trace 1: SS Trace 2: RTCT Sawtooth Trace 3: GATE Output FIGURE 11. SOFT-START WITH OVERCURRENT FAULT
Figure 12 shows the switching FET waveforms during steady state operation. Trace 1 is drain-source voltage and Trace 2 is gate-source voltage.
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Component List
REFERENCE DESIGNATOR C1, C2, C3 C5, C13 C15, C16, C19, C20 C17 C18 C21, C22 C4, C14 C6 C7 C8 C9, C10, C11, C12 CR2, CR6 CR4, CR5 D1 D2 Q1 Q2 Q3 R1, R2 R10 R7, R9, R11, R26, R27 R12 R13, R15, R17, R18, R19, R25 R14 R16 R21 R22 R24 R3, R23 R4 R5 R6 R8, R20 T1 U2 U3 U4 VR1 1.00k 20.0k 10.0k 38.3k 1.00k 10 165 10.0 5.11 3.92k 100 1.00 221k 75.0k 330pF 0.22F VALUE 1.0F 0.1F 560F 470pF 0.01F 22F 100pF 1500pF Capacitor, 1812, X7R, 100V, 20% Capacitor, 0603, X7R, 25V, 10% Capacitor, Radial, SANYO 4SEP560M Capacitor, 0603, COG, 50V, 5% Capacitor, 0805, X7R, 50V, 10% Capacitor, 1210, X5R, 10V, 20% Capacitor, 0603, COG, 50V, 5% Capacitor, Disc, Murata DE1E3KX152MA5BA01 0 Jumper, 0603 Capacitor, 0603, COG, 50V, 5% Capacitor, 0603, X7R, 16V, 10% Diode, Fairchild ES1C Diode, IR 12CWQ03FN Zener, 18V, Zetex BZX84C18 Diode, Schottky, BAT54C FET, Fairchild FDS2570 Transistor, Zetex FMMT491A Transistor, ON MJD31C Resistor, 1206, 1% Resistor, 0603, 1% Resistor, 0603, 1% Resistor, 0603, 1% Resistor, 0603, 1% Resistor, 0603, 1% Resistor, 0603, 1% Resistor, 1206, 1% Resistor, 0603, 1% Resistor, 2512, 1% Resistor, 0603, 1% Resistor, 2512, 1% Resistor, 0603, 1% Resistor, 0603, 1% OMIT Transformer, MIDCOM 31555 Opto-coupler, NEC PS2801-1 Shunt Reference, National LM431BIM3 PWM, Intersil ISL6721IB Zener, 15V, Zetex BZX84C15 DESCRIPTION
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References
1. Ridley, R., "A New Continuous-Time Model for Current Mode Control", IEEE Transactions on Power Electronics, Vol. 6, No. 2, April 1991. 2. Dixon, Lloyd H., "Closing the Feedback Loop", Unitrode Power Supply Design Seminar, SEM-700, 1990.
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FN9110.6 March 5, 2008
ISL6721 Thin Shrink Small Outline Plastic Packages (TSSOP)
N INDEX AREA E E1 -B1 2 3 L 0.05(0.002) -AD -CSEATING PLANE A 0.25 0.010 GAUGE PLANE 0.25(0.010) M BM
M16.173
16 LEAD THIN SHRINK SMALL OUTLINE PLASTIC PACKAGE INCHES SYMBOL A A1 A2 b c D MIN 0.002 0.033 0.0075 0.0035 0.193 0.169 0.246 0.020 16 0o 8o 0o MAX 0.043 0.006 0.037 0.012 0.008 0.201 0.177 0.256 0.028 MILLIMETERS MIN 0.05 0.85 0.19 0.09 4.90 4.30 6.25 0.50 16 8o MAX 1.10 0.15 0.95 0.30 0.20 5.10 4.50 6.50 0.70 NOTES 9 3 4 6 7 Rev. 1 2/02
A1 0.10(0.004) A2 c
E1 e E L N
e
b 0.10(0.004) M C AM BS
0.026 BSC
0.65 BSC
NOTES: 1. These package dimensions are within allowable dimensions of JEDEC MO-153-AB, Issue E. 2. Dimensioning and tolerancing per ANSI Y14.5M-1982. 3. Dimension "D" does not include mold flash, protrusions or gate burrs. Mold flash, protrusion and gate burrs shall not exceed 0.15mm (0.006 inch) per side. 4. Dimension "E1" does not include interlead flash or protrusions. Interlead flash and protrusions shall not exceed 0.15mm (0.006 inch) per side. 5. The chamfer on the body is optional. If it is not present, a visual index feature must be located within the crosshatched area. 6. "L" is the length of terminal for soldering to a substrate. 7. "N" is the number of terminal positions. 8. Terminal numbers are shown for reference only. 9. Dimension "b" does not include dambar protrusion. Allowable dambar protrusion shall be 0.08mm (0.003 inch) total in excess of "b" dimension at maximum material condition. Minimum space between protrusion and adjacent lead is 0.07mm (0.0027 inch). 10. Controlling dimension: MILLIMETER. Converted inch dimensions are not necessarily exact. (Angles in degrees)
21
FN9110.6 March 5, 2008
ISL6721 Small Outline Plastic Packages (SOIC)
N INDEX AREA E -B1 2 3 SEATING PLANE -AD -CA h x 45 H 0.25(0.010) M BM
M16.15 (JEDEC MS-012-AC ISSUE C)
16 LEAD NARROW BODY SMALL OUTLINE PLASTIC PACKAGE INCHES SYMBOL A
L
MILLIMETERS MIN 1.35 0.10 0.33 0.19 9.80 3.80 MAX 1.75 0.25 0.51 0.25 10.00 4.00 NOTES 9 3 4 5 6 7 8 Rev. 1 6/05
MIN 0.0532 0.0040 0.013 0.0075 0.3859 0.1497
MAX 0.0688 0.0098 0.020 0.0098 0.3937 0.1574
A1 B C D E
A1 0.10(0.004) C
e H h L N
0.050 BSC 0.2284 0.0099 0.016 16 0 8 0.2440 0.0196 0.050
1.27 BSC 5.80 0.25 0.40 16 0 6.20 0.50 1.27
e
B 0.25(0.010) M C AM BS
NOTES: 1. Symbols are defined in the "MO Series Symbol List" in Section 2.2 of Publication Number 95. 2. Dimensioning and tolerancing per ANSI Y14.5M-1982. 3. Dimension "D" does not include mold flash, protrusions or gate burrs. Mold flash, protrusion and gate burrs shall not exceed 0.15mm (0.006 inch) per side. 4. Dimension "E" does not include interlead flash or protrusions. Interlead flash and protrusions shall not exceed 0.25mm (0.010 inch) per side. 5. The chamfer on the body is optional. If it is not present, a visual index feature must be located within the crosshatched area. 6. "L" is the length of terminal for soldering to a substrate. 7. "N" is the number of terminal positions. 8. Terminal numbers are shown for reference only. 9. The lead width "B", as measured 0.36mm (0.014 inch) or greater above the seating plane, shall not exceed a maximum value of 0.61mm (0.024 inch). 10. Controlling dimension: MILLIMETER. Converted inch dimensions are not necessarily exact.
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Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see www.intersil.com 22
FN9110.6 March 5, 2008

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